Bidimensional phase-varying metamaterial for steering beam antenna

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Bidimensional Phase-varying Metamaterial for steering beam antenna

Abdelwaheb Ourir, Shah Nawaz Burokur*, André de Lustrac

Institut d’Electronique Fondamentale, Univ. Paris-Sud, UMR 8622 ; CNRS, Orsay, F-91405

ABSTRACT

Dielectric substrates supporting planar periodic subwavelength metamaterial-based metallic arrays and presenting frequency dispersive phase characteristics are applied to ultra-compact high-gain and high-directivity planar antennas. In this paper, different models of metamaterial-based surfaces introducing a zero degree reflection phase shift to incident waves are firstly studied numerically using finite-element method analysis where the bandwidth and operation frequency are predicted. These surfaces are then applied in a resonant Fabry-Perot type cavity and a ray optics analysis is used to design different models of ultra-compact high-gain microstrip printed antennas. Firstly, a cavity antenna of thickness

λ/60 based on the use of a microstrip patch antenna and two bidimensional metamaterial-based surfaces, the first one acting as a High Impedance Surface (HIS) and the second one acting as a Partially Reflecting Surface (PRS) is designed. This cavity is then optimized for easier fabrication process and loss reduction by the use of only one bidimensionnal composite metamaterial-based surface acting as a PRS. Secondly, another surface presenting a variable phase by the use of a non periodic metamaterial-based metallic strips array is designed for a passive low-profile steering beam antenna application. Finally, a switchable operation frequency cavity by the implementation of varicap diodes is designed and fabricated. All these cavity antennas operate on subwavelength modes, the smallest cavity thickness being of the order of

λ/60.

Keywords: metamaterial, high-gain antennas, ultra-compact antennas, Fabry-Perot cavity, High Impedance Surface, Partially Reflecting Surface, variable phase, steering beam antenna, switchable operation frequency, varicap diodes.

1. INTRODUCTION

There has been a lot of study published in literature on the improvement of the performances of microstrip patch antennas [1-3]. Most of the solutions proposed in the past were to use an array of several antennas. The particular disadvantage of this method comes from the feeding of each antenna and also from the coupling between each element. Other interesting solutions have been suggested: the first one was to make use of a superstrate of either high permittivity or permeability above the patch antenna [1] and the second one proposed recently by Nakano et al., is to sandwich the antenna by dielectric layers of the same permittivity [2]. The numerical study of a patch antenna where a Left-Handed Medium (LHM) is placed above has also been done and in this case a gain enhancement of about 3dB has been observed [3]. However, these solutions are all based on non-planar designs which are bulky for novel telecommunication systems requiring compact low-profile and environment friendly directive antennas.

Concerning the design of compact directive electromagnetic sources based on a single feeding point, different interesting solutions have also been proposed recently. At first, resonant cavities in one-dimensional (1-D) dielectric photonic crystals were used [4]. Afterwards, three dimensional (3-D) structures rather than 1-D were used, leading to better performances [5]. Another interesting solution proposed by Enoch et al. is to use the refractive properties of a low optical index material interface in order to achieve the directive emission [6]. We shall note that these solutions are all also based on the use of a bulky material. Otherwise, the most common method to reach directive emission is obviously based on the Fabry-Perot cavity. Such cavities can be considered quite bulky too since a thickness of half of the working wavelength is required [7]. But recently, Feresidis et al. showed that the half wavelength restriction in a Fabry-Perot cavity can be reduced to a quarter wavelength by using a novel type of metamaterial-based resonant cavity together with a microstrip patch antenna [8]. The latter requires the use of Artificial Magnetic Conductor (AMC) surfaces introducing a zero degree reflection phase shift to incident waves. The AMC surfaces have been first proposed by Sievenpiper et al. in order to act as the so called High Impedance Surface (HIS) [9]. This HIS is composed of metallic patches periodically

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organized on a dielectric substrate and shorted to the metallic ground plane with vias, appearing as “mushroom” structures. In a particular narrow frequency band, this surface creates image currents and reflections in-phase with the emitting source instead of out of phase reflections as the case of conventional metallic ground plane. The HIS allows also the suppression of surface waves which travel on conventional ground plane. New applications of improved performances in the field of reflectors and antennas have then been proposed. However, the HIS of Sievenpiper needs a non-planar fabrication process, which is not suitable for implementation in lots of microwave and millimetric circuits. Other models of planer AMC surfaces have then been proposed for antenna and circuit applications [10, 11]. The cavity proposed by Feresidis is composed of two planar AMC surfaces. The first one is used as a HIS so as to replace the Perfect Electric Conductor (PEC) surface for the antenna and hence, to avoid the λ/4 limit distance between the source and the ground plane. The second one acts as a Partially Reflective Surface (PRS) with a reflection phase equal to 180°. This idea has been pushed further by Zhou et al. by taking advantage of the dispersive characteristics of metamaterials, designing a subwavelength cavity with a thickness smaller than a 10th of the wavelength [12]. Compared to Feresidis, Zhou made use of a non-planar mushroom structure with a dipole acting as the feeding source.

In this paper, using a novel composite metamaterial, made of capacitive and inductive grids, we review our works in the fields of low-profile and high-gain metamaterial-based cavity antennas. We will show how our group has lately further reduced the cavity thickness by λ/60 for applications to ultra-thin directive antennas using two bidimensional metamaterial-based surfaces, one as a HIS and the other as a PRS [13]. An optimization of the cavity has also been undertaken in order to facilitate the fabrication process and also to reduce the metallic losses by using a PEC surface as the antenna’s ground plane and only one subwavelength metamaterial-based composite surface as the PRS [14]. We will then present the modeling and characterization of an optimized resonant cavity for a reconfigurable directive beam antenna near 10 GHz. The cavity is composed of a PEC surface and a new unidimensional composite metamaterial made of metallic strips composing a non uniform capacitive grid and a uniform inductive grid, acting as the PRS [15]. Finally, we will present our first results obtained with an electronically active metamaterial-based subwavelength cavity for a frequency reconfigurable low-profile and high-gain antenna application. A numerical analysis using the finite-element method software HFSS [16] together with discussions on the fabrication process and the characterization results will be presented for the cavities mentioned above.

2. ANALYSIS OF THE PLANAR SUBWAVELENGTH METAMATERIAL-BASED

SURFACES

A cavity antenna is formed by a feeding source placed between two reflecting surfaces. In this paper, the two models of cavities presented in Fig.1 will be discussed and used. The cavity depictured in Fig. 1(a) is composed of a PEC surface acting as a conventional ground plane for the feeding source and an metamaterial-based surface playing the role of a transmitting window known as PRS. The other one presented in Fig. 1(b) is formed by two metamaterial-based surfaces, one playing the role of a HIS for the feeding source and the other one a PRS.

(a) (b)

Fig. 1. Resonant cavity formed by (a) a PEC and a PRS, (b) two metamaterial-based surfaces: a HIS and a PRS, with a microstrip antenna as the feeding source inside the cavity.

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Whichever cavity we are going to use, it requires the application of a metamaterial-based surface. So in this section, we will design planar metamaterial-based surfaces for operations near 10 GHz. The surface used by our group in order to achieve the HIS is made of a metamaterial composed of 2-D periodically subwavelength metallic square patches organized on one face of a dielectric substrate as illustrated in Fig. 2(a). The different dimensions of the patches are as follows: period p1 = 4 mm and width w1 = 3.8 mm. Another surface which we are going to use for the PRS of the cavity is made of a composite metamaterial consisting of simultaneously a capacitive and an inductive grid on the two faces of a dielectric substrate. The capacitive grid is also formed by 2-D periodic metallic patches (period p2 = 4 mm and width

w2 = 3.6 mm) whereas the inductive grid is formed by a 2-D periodic mesh (line width l = 1.2 mm) as shown in Fig. 2(b). Concerning the substrate, we have used the double copper cladded FR3 epoxy of relative permittivity εr = 3.9 and having a thickness of 1.4 mm. The size of the different patterns has been chosen in order to minimize the phase of the reflection coefficient near 10 GHz while providing a sufficiently high reflectance (~90%).

(a) (b)

Fig. 2. Metamaterial-based surface: (a) HIS formed by periodic metallic patches, (b) PRS composed of a capacitive and inductive grid respectively formed by periodic metallic patches and a periodic mesh.

Following the early work of Trentini [17], recently revisited by Feresidis et al. [8], a simple optical ray model can be used to describe the resonant cavity modes. This model is used to theoretically predict the existence of a low-profile high-directivity metamaterial-based subwavelength cavity antenna. Let us consider the cavities presented in Fig. 1. They are formed by a feeding antenna placed between two reflectors separated by a distance h. Phase shifts are introduced by these two reflectors and also by the path length of the wave travelling inside the cavity. With the multiple reflections of the wave emitted by the antenna, a resonance is achieved when the reflected waves are in phase after one cavity roundtrip. The resonance condition can then be written as:

h =

(

)

2 N

4

r PRS

λ

π

λ

φ

+

±

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where φPRS is the reflection phase of the PRS reflector, φr is the reflection phase of the reflector near the antenna (either PEC or HIS reflector), and Nis an integer. If the cavity thickness h is fixed, the resonant wavelength is determined by the sum of the reflection phases φPRS + φr. Conversely, for a given wavelength, the thickness h can be minimized by minimizing the total phase shift φPRS + φr. The use of metamaterial-based surfaces answers this purpose since they provide a low reflection phase around the frequency of maximum surface impedance. If the reflector near the feeding antenna is composed of a PEC surface, then φr will be very close to 180°. In the case of a HIS, φr will be equal to 0° and situations can occur where the reflection phases of the two metamaterial-based surfaces have different signs and the sum

φPRS + φr is close to zero.

The two metamaterial-based surfaces of the HIS-PRS cavity antenna are analyzed numerically using the finite-element software HFSS so as to predict the evolution of the cavity thickness h versus frequency. The simulations are performed on one unit cell together with appropriate boundary conditions. The results are presented in Fig. 3. As shown, the calculated resonance frequency of the HIS and PRS reflectors are respectively 10.4 GHz and 9.7 GHz (Fig. 3(a) and 3(b)). These resonance frequencies can also be seen on the traces of Fig. 3(c) when the reflection phase crosses 0°. These two surfaces are fabricated on the FR3 epoxy substrate by a mechanical milling process using an LPKF Protomat40 machine. The reflection and transmission measurements are performed by using two microwave horn

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antennas connected to an Agilent 8722ES network analyzer. The results issued from measurements shown in Fig. 3(a), 3(b) and 3(c) agree very well with the calculated results. The different phases (simulated and measured) are used to estimate the thickness h of the HIS-PRS cavity as given by Eq. (1). Fig. 3(d) shows that h first decreases with increasing frequency of the first resonant mode (N = 0) to the point where a cavity zero thickness is reached at around 10.2 GHz. Then a jump in the mode occurs leading to an abrupt variation of h and the value decreases again for N = 1.

(a) _(b)

Fig. 3. (a) Calculated and measured reflection magnitude for the HIS reflector. (b) Calculated and measured reflection magnitude for the PRS reflector. (c) Calculated and measured reflection phases for the PRS and HIS reflectors. (d) Evolution of the cavity thickness h versus frequency, this evolution being estimated from Eq. (1) by the calculated and measured reflection phases reported in (c).

3. METAMATERIAL-BASED HIGH-DIRECTIVITY LOW-PROFILE CAVITY

ANTENNA

3.1 HIS-PRS cavity antenna

The HIS-PRS cavity antenna is formed by the two metamaterial-based reflectors (HIS and PRS) mentioned in the previous section together with a patch antenna designed to operate at about 10 GHz. The patch antenna of dimensions 6.8 × 7 mm2 is placed on the HIS in the cavity as shown in Fig. 4(a). This figure also illustrates the experimental setup for the measurements of the antenna radiation patterns in an anechoic chamber. A cavity thickness h = 0.5 mm is chosen

N=0

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for the cavity. The thickness h of the Fabry-Perot cavity formed by the two reflectors is adjusted mechanically. The lateral dimensions of the reflector plates are 17 × 17 cm2. This thickness leads to a good matching of the cavity at 10.1 GHz (Fig. 4(b)) corresponding to the design of a λ/60 cavity. This frequency is in good agreement with the resonance frequency calculated from the optical ray model. The directive emission of the subwavelength cavity antenna at 10.1 GHz is illustrated from the calculated and measured E-plane (φ = 90°) and H-plane (φ = 0°) radiation patterns in Fig. 4(c) and 4(d).

(a) (b)

Fig. 4. (a) Schematic view of the experimental setup for measuring the radiation patterns of the cavity antenna. The cavity is formed by two metamaterial-based reflectors (HIS and PRS) together with a patch antenna placed on the HIS. (b) Calculated and measured reflection magnitude for the PRS reflector. (c) E-plane (φ = 90°) radiation pattern of the HIS-PRS cavity antenna at 10.1 GHz. (d) H-plane (φ = 0°) radiation pattern of the HIS-PRS cavity antenna at 10.1 GHz.

Using the formulation proposed in [5], the directivity of the cavity antenna is written as: D =

2 1

4 θ

π

(1)

where θ1 and θ2 are respectively the half-power widths for the E-plane and H-plane radiation patterns. The antenna directivity is then found to be equal to 78 (19 dB) for θ1 = 22° and θ2 = 24°.

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3.2 PEC-PRS cavity antenna

In order to simplify the fabrication of the cavity antenna, another one using only one metamaterial-based surface reflector acting as the PRS and a PEC reflector (similar to the cavity shown in Fig. 1(a)) is designed. As we have seen from the reflection coefficients in Fig. 3(a) and 3(b), losses are maximum at the resonance frequency of the metamaterial-based surfaces. Thus using only one reflector has also the advantage of presenting lower losses. A new PRS composed of simultaneously a capacitive and an inductive grid on the two faces of a dielectric substrate is designed (Fig. 2(b)). Concerning the metallic patches of the capacitive grid, a period p2 = 5 mm and a width w2 = 4.8 mm are used. A line width l = 2.2 mm is considered for the mesh of the inductive grid. This PRS having a resonance frequency of about 8 GHz presents a reflection phase close to -150° for frequencies higher than 10 GHz (Fig. 5 (a)) . The use of such a reflector in conjunction with a PEC leads also to a subwavelength cavity since the sum (φPRS + φr) is very close to zero between 9 GHz and 11 GHz.

A 1 mm (λ/30) thick cavity is designed with lateral dimensions of 10 × 10 cm2 where the resonance is achieved at around 9.7 GHz. At this resonance frequency, the cavity is well matched. The antenna gain patterns in the E- and H-planes obtained from simulation and measurements are presented in Fig. 5 (b) and 5(c).

(a) (b) (c)

Fig. 5. (a) Calculated and measured reflection phase for the PRS reflector. Radiation patterns of the cavity antenna formed by one metamaterial-based reflector (PRS) and a PEC reflector together with a patch antenna at 9.7 GHz: (b) in the E-plane (φ = 90°), (c) in the H-plane (φ = 0°).

In this case, despite the use of only one metamaterial-based surface as the PRS and the use of smaller lateral dimensions than the two metamaterial-based cavity, the antenna directivity is found to be twice and equal to 160 (22 dB).

4. PASSIVE STEERING BEAM CAVITY ANTENNA

In this section, we report the design of a steerable directive beam antenna using a resonant metamaterial-based subwavelength cavity (Fig. 6(a)). Generally, phased-arrayed antennas are used by imposing electrical phase differences between the antenna elements so as to steer a beam into a desired direction [18]. Although this solution offers fast scan speed, it requires many phase shifters connected to each antenna element, which is very expensive for communication systems. Our approach here is based on a cavity made of a conventional PEC ground plane and a novel composite unidimensional metamaterial with a locally variable phase acting as a Partially Reflective Surface (PEC-PRS cavity of Fig. 1(a)). The antenna used is a rectangular microstrip patch designed to operate near 10 GHz. So, first of all we will describe the study and design of this new phase-varying metamaterial and then its application for a cavity antenna.

4.1 Composite unidimensional metamaterial with locally variable phase

The PRS reflector used for the modelling of the steerable directive beam antenna consists of a periodic array of copper strips mechanically etched on each face of the 1.4 mm thick FR3-epoxy (εr = 3.9 and tan( ) = 0.0197) substrate as shown in Fig. 6(b). The periodicity a and the width w of the strips are respectively 5 mm and 1.2 mm, optimized to have a

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resonance near 10 GHz and to provide a sufficiently high reflectance. The upper array where the strips are oriented parallel to the electric field E of the antenna plays the role of the inductive grid, whereas the lower array where the strips are oriented parallel to the magnetic field H acts as the capacitive grid. By changing the gap spacing g between the metallic strips of the capacitive grid of the PRS in only the direction parallel to E (x–direction is chosen here) and keeping all the other geometric parameters unchanged, the capacitance of the metamaterial will also vary along this same direction. As a consequence, the phases of the computed reflection and transmission coefficients vary. This behaviour is illustrated by the simulation results of the transmission coefficient phase shown in Fig. 7(a). Due to the different gap spacing g, each PRS has a different transmission phase characteristic. We can note that the variation of g accounts for the shift of the resonance frequency. An increase in the value of g causes a decrease in the value of the capacitance created between two cells, and finally a shift of the resonance towards higher frequencies. At a particular frequency the phase of the PRS increases with an increase in the gap spacing. This phase shift especially for the transmission coefficient is very important since it will help controlling the radiated beam direction of the antenna.

(a) (b)

Fig. 6. (a) Schematic view of the cavity composed of a PEC and a metamaterial-based phase-varying PRS. (b) Elementary cell of the composite metamaterial, showing the capacitive and inductive grids.

4.2 Steerable directive antenna

In order to make the antenna beam steerable, the PRS of the cavity must be designed in such a manner that it exists a continuous phase variation. The precedent study on the variation of g leads us to a new design of the PRS (Fig. 6) with a continuous variation of the gap g. As illustrated by the cavity antenna system shown in Fig. 6(a), the proposed PRS is now divided into seven different regions, where each of them has a specific gap spacing. The basic gap spacing g (g = 400 µm) in the centre of the PRS is taken as a reference and a double variation of ± n g (with n = 0, 1, 2, 3) is imposed on each side of the central region. The gap spacing is thus incremented regularly when moving from the left to the right along the PRS. If we consider each gap as a slot antenna, an analogy can then be made with an array of seven antennas with a regular phase difference. Note that here the resonance frequency of the central region of the metamaterial corresponds to that of the PRS without gap spacing variation (g = 400 µm and g = 0), i.e. 8.7 GHz as shown in Fig. 7(a).

Several prototypes of the subwavelength cavity have been simulated and fabricated. The first one consists of the metamaterial PRS with the same gap spacing g between the metallic strips of the capacitive grid ( g = 0). This prototype will assure no deflection of the beam since it exists no phase variation of the metamaterial. The second and third one are the prototypes incorporating respectively a variation of g = 50 µm and g = 100 µm along the positive x-direction. The cases where the variation g is negative (180° rotation of the PRS around the z-axis) have also been considered. The resonance frequency of the proposed cavity which depends on the phase of the reflection coefficient of the PRS and also the height is found to be 10.5 GHz. Fig. 7(b) show the measured gain patterns of the antenna in the E (Ø = 90°) plane at 10.5 GHz for an optimized cavity thickness h = 1 mm, a = 5 mm, w = 2.2 mm, g = 400 µm, g = 0. The beam is normal to the plane of the antenna and shows no deflection, which confirms our prediction on the constant phase metamaterial. However, in the case of a regular variation of 50 µm, a deflection of the antenna beam of about 10° can be observed either in the forward (clockwise) or backward (anti-clockwise) direction depending if g is respectively negative or positive. Similar observations and a higher deflection of ±20° can be noted for g = ±100 µm. This figure illustrates very clearly the control of the direction of the radiation beam when the gap spacing, and consequently the phase, vary locally along the PRS surface. However, a side-lobe, which is mainly due to the asymmetrical feeding of the microstrip antenna, appears on the gain pattern for the case g = –100 µm.

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(a) (b)

Fig. 7. (a) Transmission coefficient phase of the metamaterial-based PRS (shown in inset) with different gap spacing g. (b) Measured gain patterns of the cavity antenna in the E-plane (φ = 90°) at 10.5 GHz for g = ±100 µm, g = ±50 µm and

g = 0 µm.

5. ELECTRONICALLY RECONFIGURABLE CAVITY ANTENNA

This section deals with the modeling and characterization of an active reconfigurable subwavelength metamaterial-based cavity antenna. The aim of this active reconfigurable cavity is to be able to control dynamically the antenna beam in the near future. Electronically controllable textured surfaces have been applied to leaky wave antennas [19] and radomes [20]. The tunable PRS used in this section is composed of a composite phase varying metamaterial, by the insertion of active electronic components, and is proposed for the design of the reconfigurable Fabry-Perot cavity antenna. The study and design of this active phase-varying metamaterial tunable PRS is described and its application for a frequency switching cavity antenna is presented.

5.1 Electronically phase-varying metamaterial

This PRS is the same as the one as illustrated in Fig. 6(b). Instead of applying a linear variation of the gap spacing g in order to create a locally variable phase, we now use active components to make the phase of the PRS shift in frequency. Varicap diodes are thus incorporated into the capacitive grid between two adjacent metallic strips (Fig. 8(a)) and depending on the applied bias voltage, the phase of the PRS varies with frequency. A prototype is designed and fabricated where all the gaps have the same capacitance according to the bias voltage applied. The variable capacitive grid of the tunable phase PRS used for this work consists of a lattice of metallic strips with varicap diodes connected each 6 mm (s = 6 mm) between two adjacent strips. The width of the strips and the spacing between two strips of the capacitive grid is respectively w = 1 mm and g = 2 mm (Fig. 8(b)). Concerning the inductive grid, the width of the strips and the spacing between two strips are respectively w1 = 2 mm and g1 = 4 mm (Fig. 8(c)). Note that the inductive grid is not made tunable.

Measurements are done in an anechoic chamber so as to observe the frequency shift of the reflection phase or also the phase jump at a desired frequency when the bias voltage is tuned. Fig. 9(a) shows the measured reflection phase coefficient of the PRS for a bias voltage of 0, 2 and 5 V. When the bias voltage is tuned from 0 to 5 V, the transmission phase response shifts towards higher frequencies. Particularly the resonance frequency of the PRS, corresponding to the null of the phase, shifts from 7.5 to 7.9 GHz. This is explained by the fact that when the bias voltage increases, the capacitance of the PRS decreases causing a shift towards higher frequencies.

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(a)

(b) (c)

Fig. 8. (a) Structure of the cavity antenna. (b) Photo of the prototype showing the side of capacitive grid of the PRS with the varicap diodes. (c) Inductive grid of the PRS.

(a) (b)

Fig. 9. (a) Electronically phase-varying PRS. (a) Reflection phase coefficient (S11 (deg)) for 0 V, 2 V and 5 V. (b) Measured

return loss (S11 (dB)) of the cavity for 0 to 10 V applied bias voltage.

5.2 Reconfigurable cavity antenna

The active PRS is then implemented in an ultra-compact cavity of thickness h = 0.5 mm with a circular patch antenna of radius 5.4 mm operating at 8 GHz as the feeding source. By tuning the bias voltage from 0 to 10 V, two important observations are made and are illustrated in Fig. 9 (b). Firstly, as for the phase response, a shift towards high frequencies of the cavity resonance is noted when the bias voltage varies from 0 V to 10 V. This is due to the decrease in the capacitance of the PRS. Moreover, it can be observed that the dip of the resonance increases when the bias voltage is tuned from 0 to 10 V, indicating an enhancement of the matching of the cavity. This enhancement is explained by the

h

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fact that the cavity antenna is better and better matched for the thickness h = 0.5 mm fixed when the bias voltage varies from 0 V to 10 V. If we change the thickness h of the cavity, we obtain a better matching at another frequency since the resonance frequency depends on the reflection phase of the PRS as well as the cavity thickness h as shown in the relation given by Eq. (1).

Measurements of the radiation patterns of the reconfigurable cavity antenna are done in the anechoic chamber and are presented in Fig. 10. These patterns in the E-plane (Fig. 10(a)) and H-plane (Fig. 10(b)) are plotted for the case of the microstrip patch antenna alone and for the cavity antenna at different frequencies corresponding to the best matching obtained when the bias voltage is at 1 V, 3 V, 6 V and 10 V. When the capacitance of the PRS changes by a variation of the bias voltage of the varicap diodes, a better directivity is obtained for the metamaterial-based cavity antenna compared to the case of the antenna alone. A -3dB beamwidth of 50° is thus observed for the cavity antenna instead of 100° for the microstrip antenna (Fig. 10(b)). However, we can note from Fig. 10(a) that a secondary lobe with quite a high magnitude appears on the radiation pattern. This high magnitude may be probably due to a high coupling between the metamaterial-based PRS and the feeding antenna. An optimization of this cavity should reduce the level of this secondary lobe and enhance the performances.

(a) (b)

Fig. 10. Radiation patterns of the patch antenna alone and the electronically reconfigurable cavity antenna with a bias voltage of 1 V, 3 V, 6 V and 10 V: (a) E-plane (φ = 90°), (b) H-plane (φ = 0°).

6. CONCLUSION

Subwavelength metamaterial-based planar surfaces are used for applications to ultra-compact highly directive cavity antennas. Two types of metamaterial-based cavities are proposed. The first one consists of using two metamaterial-based surfaces; one acting as a High Impedance Surface (HIS) composed of periodic square metallic patches and another one acting as a Partially Reflecting Surface composed of periodic metallic patches and a mesh printed on the two faces of a dielectric substrate. The second cavity uses only the metamaterial-based surface as the PRS together with the Perfect Electric Conductor (PEC) served as the feeding antenna’s ground plane. With the first HIS-PRS cavity, a directivity of 78 (19dB) is obtained for a cavity thickness of 0.5 mm (λ/60 at 10 GHz). The PEC-PRS cavity presents a higher directivity of 160 (22dB) for a cavity thickness of 1 mm (λ/30). A new locally phase-varying metamaterial-based PRS is also proposed for applications to a cavity antenna where the radiated beam can be steered into a desired direction. This phase-varying PRS is composed of inductive and capacitive strips and by varying the capacitance locally with a continuous variation of the gap between the capacitive strips, a variable phase is achieved. A fabricated compact steerable directive cavity antenna of thickness λ/30 shows the possibility of obtaining a deflection of the beam of ±20°. Finally, an electronically tunable PRS composed of a composite phase varying metamaterial, by the incorporation of active electronic components (varicap diodes), is also proposed for the design of a frequency reconfigurable cavity antenna. A shift in the resonance frequency of the cavity antenna is noted when the bias voltage of the varicap diodes

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varies. Additional work is required for the optimization of this electronically tunable PEC-PRS cavity antenna in order to reduce the level of the secondary lobe.

For the future, we intend to design an electronically phase-varying PRS so as to implement it into a steerable ultra-compact cavity antenna.

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15. A. Ourir, S. N. Burokur, and A. de Lustrac, “Steerable ultra-thin directive antenna from a metamaterial-based subwavelength cavity,” in Proc. International Workshop on Antenna Technology 2007, Cambridge, UK (2007). 16. HFSS, High-frequency structure simulator ver. 10.1, Finite-element package, Ansoft Corporation, Pittsburgh, PA,

2006.

17. G. V. Trentini, “Partially reflecting sheet arrays,” IRE Trans. Antennas Propag. 4, 666-671 (1956). 18. C. A. Balanis, Antenna Theory: Analysis and Design (2nd edition), Wiley-Interscience, New York (1996).

19. D. F. Sievenpiper, “Forward and backward leaky wave radiation with large effective aperture from an electronically tunable textured surface,” IEEE Trans. Antennas Propag. 53(1), 236-247 (2005).

20. H. Chen B.-I. Wu, L. Ran, T. M. Grzegorczyk, and J. A. Kong, “Controllable left-handed metamaterial and its application to a steerable antenna,” Appl. Phys. Lett. 89, 053509-1– 053509-3 (2006).

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3.2 PEC-PRS cavity antenna

A line width l = 2.2 mm is considered for the mesh of the inductive grid. This PRS having a resonance frequency of about 8 GHz presents a reflection phase close to -150° for frequencies higher than 10 GHz (Fig. 5 (a)) . The use of such a reflector in conjunction with a PEC leads also to a subwavelength cavity since the sum (φPRS + φr) is very close to zero

between 9 GHz and 11 GHz.

(a) (b) (c)

4. PASSIVE STEERING BEAM CAVITY ANTENNA

4.1 Composite unidimensional metamaterial with locally variable phase

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resonance near 10 GHz and to provide a sufficiently high reflectance. The upper array where the strips are oriented parallel to the electric field E of the antenna plays the role of the inductive grid, whereas the lower array where the strips are oriented parallel to the magnetic field H acts as the capacitive grid. By changing the gap spacing g between the metallic strips of the capacitive grid of the PRS in only the direction parallel to E (x–direction is chosen here) and keeping all the other geometric parameters unchanged, the capacitance of the metamaterial will also vary along this same direction. As a consequence, the phases of the computed reflection and transmission coefficients vary. This behaviour is illustrated by the simulation results of the transmission coefficient phase shown in Fig. 7(a). Due to the different gap spacing g, each PRS has a different transmission phase characteristic. We can note that the variation of g accounts for the shift of the resonance frequency. An increase in the value of g causes a decrease in the value of the capacitance created between two cells, and finally a shift of the resonance towards higher frequencies. At a particular frequency the phase of the PRS increases with an increase in the gap spacing. This phase shift especially for the transmission coefficient is very important since it will help controlling the radiated beam direction of the antenna.

(a) (b)

Fig. 6. (a) Schematic view of the cavity composed of a PEC and a metamaterial-based phase-varying PRS. (b) Elementary cell of the composite metamaterial, showing the capacitive and inductive grids.

4.2 Steerable directive antenna

In order to make the antenna beam steerable, the PRS of the cavity must be designed in such a manner that it exists a continuous phase variation. The precedent study on the variation of g leads us to a new design of the PRS (Fig. 6) with a continuous variation of the gap g. As illustrated by the cavity antenna system shown in Fig. 6(a), the proposed PRS is now divided into seven different regions, where each of them has a specific gap spacing. The basic gap spacing g (g = 400 µm) in the centre of the PRS is taken as a reference and a double variation of ± n g (with n = 0, 1, 2, 3) is imposed on each side of the central region. The gap spacing is thus incremented regularly when moving from the left to the right along the PRS. If we consider each gap as a slot antenna, an analogy can then be made with an array of seven antennas with a regular phase difference. Note that here the resonance frequency of the central region of the metamaterial corresponds to that of the PRS without gap spacing variation (g = 400 µm and g = 0), i.e. 8.7 GHz as shown in Fig. 7(a).

Several prototypes of the subwavelength cavity have been simulated and fabricated. The first one consists of the metamaterial PRS with the same gap spacing g between the metallic strips of the capacitive grid ( g = 0). This prototype will assure no deflection of the beam since it exists no phase variation of the metamaterial. The second and third one are the prototypes incorporating respectively a variation of g = 50 µm and g = 100 µm along the positive x-direction. The cases where the variation g is negative (180° rotation of the PRS around the z-axis) have also been considered. The resonance frequency of the proposed cavity which depends on the phase of the reflection coefficient of the PRS and also the height is found to be 10.5 GHz. Fig. 7(b) show the measured gain patterns of the antenna in the E (Ø = 90°) plane at 10.5 GHz for an optimized cavity thickness h = 1 mm, a = 5 mm, w = 2.2 mm, g = 400 µm, g = 0. The beam is normal to the plane of the antenna and shows no deflection, which confirms our prediction on the constant phase metamaterial. However, in the case of a regular variation of 50 µm, a deflection of the antenna beam of about 10° can be observed either in the forward (clockwise) or backward (anti-clockwise) direction depending if g is respectively negative or positive. Similar observations and a higher deflection of ±20° can be noted for g = ±100 µm. This figure illustrates very clearly the control of the direction of the radiation beam when the gap spacing, and consequently the phase, vary locally along the PRS surface. However, a side-lobe, which is mainly due to the asymmetrical feeding of the microstrip antenna, appears on the gain pattern for the case g = –100 µm.

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(a) (b)

g = 0 µm.

5. ELECTRONICALLY RECONFIGURABLE CAVITY ANTENNA

5.1 Electronically phase-varying metamaterial

not made tunable.

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(a)

(b) (c)

Fig. 8. (a) Structure of the cavity antenna. (b) Photo of the prototype showing the side of capacitive grid of the PRS with the varicap diodes. (c) Inductive grid of the PRS.

(a) (b)

Fig. 9. (a) Electronically phase-varying PRS. (a) Reflection phase coefficient (S11 (deg)) for 0 V, 2 V and 5 V. (b) Measured return loss (S11 (dB)) of the cavity for 0 to 10 V applied bias voltage.

5.2 Reconfigurable cavity antenna

h

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(a) (b)

6. CONCLUSION

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varies. Additional work is required for the optimization of this electronically tunable PEC-PRS cavity antenna in order to reduce the level of the secondary lobe.

For the future, we intend to design an electronically phase-varying PRS so as to implement it into a steerable ultra-compact cavity antenna.

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15. A. Ourir, S. N. Burokur, and A. de Lustrac, “Steerable ultra-thin directive antenna from a metamaterial-based subwavelength cavity,” in Proc. International Workshop on Antenna Technology 2007, Cambridge, UK (2007). 16. HFSS, High-frequency structure simulator ver. 10.1, Finite-element package, Ansoft Corporation, Pittsburgh, PA,

2006.

17. G. V. Trentini, “Partially reflecting sheet arrays,” IRE Trans. Antennas Propag. 4, 666-671 (1956). 18. C. A. Balanis, Antenna Theory: Analysis and Design (2nd edition), Wiley-Interscience, New York (1996).

19. D. F. Sievenpiper, “Forward and backward leaky wave radiation with large effective aperture from an electronically tunable textured surface,” IEEE Trans. Antennas Propag. 53(1), 236-247 (2005).

20. H. Chen B.-I. Wu, L. Ran, T. M. Grzegorczyk, and J. A. Kong, “Controllable left-handed metamaterial and its application to a steerable antenna,” Appl. Phys. Lett. 89, 053509-1– 053509-3 (2006).

Bidimensional phase-varying metamaterial for steering beam antenna