organized on a dielectric substrate and shorted to the metallic ground plane with vias, appearing as “mushroom” structures. In a particular narrow frequency band, this surface creates image currents and reflections in-phase with the
emitting source instead of out of phase reflections as the case of conventional metallic ground plane. The HIS allows also the suppression of surface waves which travel on conventional ground plane. New applications of improved
performances in the field of reflectors and antennas have then been proposed. However, the HIS of Sievenpiper needs a non-planar fabrication process, which is not suitable for implementation in lots of microwave and millimetric circuits.
Other models of planer AMC surfaces have then been proposed for antenna and circuit applications [10, 11]. The cavity proposed by Feresidis is composed of two planar AMC surfaces. The first one is used as a HIS so as to replace the
Perfect Electric Conductor PEC surface for the antenna and hence, to avoid the
λ4 limit distance between the source and the ground plane. The second one acts as a Partially Reflective Surface PRS with a reflection phase equal to 180°.
This idea has been pushed further by Zhou et al. by taking advantage of the dispersive characteristics of metamaterials, designing a subwavelength cavity with a thickness smaller than a 10th of the wavelength [12]. Compared to Feresidis,
Zhou made use of a non-planar mushroom structure with a dipole acting as the feeding source.
In this paper, using a novel composite metamaterial, made of capacitive and inductive grids, we review our works in the fields of low-profile and high-gain metamaterial-based cavity antennas. We will show how our group has lately further
reduced the cavity thickness by λ60 for applications to ultra-thin directive antennas using two bidimensional
metamaterial-based surfaces, one as a HIS and the other as a PRS [13]. An optimization of the cavity has also been undertaken in order to facilitate the fabrication process and also to reduce the metallic losses by using a PEC surface as
the antenna’s ground plane and only one subwavelength metamaterial-based composite surface as the PRS [14]. We will then present the modeling and characterization of an optimized resonant cavity for a reconfigurable directive beam
antenna near 10 GHz. The cavity is composed of a PEC surface and a new unidimensional composite metamaterial made of metallic strips composing a non uniform capacitive grid and a uniform inductive grid, acting as the PRS [15]. Finally,
we will present our first results obtained with an electronically active metamaterial-based subwavelength cavity for a frequency reconfigurable low-profile and high-gain antenna application. A numerical analysis using the finite-element
method software HFSS [16] together with discussions on the fabrication process and the characterization results will be presented for the cavities mentioned above.
2. ANALYSIS OF THE PLANAR SUBWAVELENGTH METAMATERIAL-BASED
SURFACES
A cavity antenna is formed by a feeding source placed between two reflecting surfaces. In this paper, the two models of cavities presented in Fig.1 will be discussed and used. The cavity depictured in Fig. 1a is composed of a PEC surface
acting as a conventional ground plane for the feeding source and an metamaterial-based surface playing the role of a transmitting window known as PRS. The other one presented in Fig. 1b is formed by two metamaterial-based surfaces,
one playing the role of a HIS for the feeding source and the other one a PRS.
a b
Fig. 1. Resonant cavity formed by a a PEC and a PRS, b two metamaterial-based surfaces: a HIS and a PRS, with a microstrip antenna as the feeding source inside the cavity.
Whichever cavity we are going to use, it requires the application of a metamaterial-based surface. So in this section, we will design planar metamaterial-based surfaces for operations near 10 GHz. The surface used by our group in order to
achieve the HIS is made of a metamaterial composed of 2-D periodically subwavelength metallic square patches organized on one face of a dielectric substrate as illustrated in Fig. 2a. The different dimensions of the patches are as
follows: period p
1
= 4 mm and width w
1
= 3.8 mm. Another surface which we are going to use for the PRS of the cavity is made of a composite metamaterial consisting of simultaneously a capacitive and an inductive grid on the two faces of
a dielectric substrate. The capacitive grid is also formed by 2-D periodic metallic patches period p
2
= 4 mm and width w
2
= 3.6 mm whereas the inductive grid is formed by a 2-D periodic mesh line width l = 1.2 mm as shown in Fig. 2b. Concerning the substrate, we have used the double copper cladded FR3 epoxy of relative permittivity
ε
r
= 3.9 and having a thickness of 1.4 mm. The size of the different patterns has been chosen in order to minimize the phase of the reflection
coefficient near 10 GHz while providing a sufficiently high reflectance ~90.
a b
Fig. 2. Metamaterial-based surface: a HIS formed by periodic metallic patches, b PRS composed of a capacitive and inductive grid respectively formed by periodic metallic patches and a periodic mesh.
Following the early work of Trentini [17], recently revisited by Feresidis et al. [8], a simple optical ray model can be used to describe the resonant cavity modes. This model is used to theoretically predict the existence of a low-profile
high-directivity metamaterial-based subwavelength cavity antenna. Let us consider the cavities presented in Fig. 1. They are formed by a feeding antenna placed between two reflectors separated by a distance h. Phase shifts are introduced by
these two reflectors and also by the path length of the wave travelling inside the cavity. With the multiple reflections of the wave emitted by the antenna, a resonance is achieved when the reflected waves are in phase after one cavity
roundtrip. The resonance condition can then be written as:
h =
2 N
4
r PRS
λ π
λ φ
φ ±
+
1 where
φ
PRS
is the reflection phase of the PRS reflector, φ
r
is the reflection phase of the reflector near the antenna either PEC or HIS reflector, and N is an integer. If the cavity thickness h is fixed, the resonant wavelength is determined by
the sum of the reflection phases φ
PRS
+ φ
r
. Conversely, for a given wavelength, the thickness h can be minimized by minimizing the total phase shift
φ
PRS
+ φ
r
. The use of metamaterial-based surfaces answers this purpose since they provide a low reflection phase around the frequency of maximum surface impedance. If the reflector near the feeding
antenna is composed of a PEC surface, then φ
r
will be very close to 180°. In the case of a HIS, φ
r
will be equal to 0° and situations can occur where the reflection phases of the two metamaterial-based surfaces have different signs and the sum
φ
PRS
+ φ
r
is close to zero. The two metamaterial-based surfaces of the HIS-PRS cavity antenna are analyzed numerically using the finite-element
software HFSS so as to predict the evolution of the cavity thickness h versus frequency. The simulations are performed on one unit cell together with appropriate boundary conditions. The results are presented in Fig. 3. As shown, the
calculated resonance frequency of the HIS and PRS reflectors are respectively 10.4 GHz and 9.7 GHz Fig. 3a and 3b. These resonance frequencies can also be seen on the traces of Fig. 3c when the reflection phase crosses 0°.
These two surfaces are fabricated on the FR3 epoxy substrate by a mechanical milling process using an LPKF Protomat40 machine. The reflection and transmission measurements are performed by using two microwave horn
antennas connected to an Agilent 8722ES network analyzer. The results issued from measurements shown in Fig. 3a, 3b and 3c agree very well with the calculated results. The different phases simulated and measured are used to
estimate the thickness h of the HIS-PRS cavity as given by Eq. 1. Fig. 3d shows that h first decreases with increasing frequency of the first resonant mode N = 0 to the point where a cavity zero thickness is reached at around 10.2 GHz.
Then a jump in the mode occurs leading to an abrupt variation of h and the value decreases again for N = 1.
a b
c d
Fig. 3. a Calculated and measured reflection magnitude for the HIS reflector. b Calculated and measured reflection magnitude for the PRS reflector. c Calculated and measured reflection phases for the PRS and HIS reflectors. d
Evolution of the cavity thickness h versus frequency, this evolution being estimated from Eq. 1 by the calculated and measured reflection phases reported in c.
3. METAMATERIAL-BASED HIGH-DIRECTIVITY LOW-PROFILE CAVITY
