Transistor Base Drive Applications

3.5 Transistor Base Drive Applications

–V cc

A plethora of circuits have been suggested to successfully com- FIGURE 3.15 mand transistors for operating in power electronics switching Non-isolated base driver.

systems. Such base drive circuits try to satisfy the following requirements: supply the right collector current, adapt the base current to the collector current, and extract a reverse cur- rent from base to speed up the device blocking. A good base driver reduces the commutation times and total losses, increas- ing efficiency and operating frequency. Depending upon the

+V cc

grounding requirements between the control and the power circuits, the base drive might be isolated or non-isolated types. R 1

Fig. 3.15 shows a non-isolated circuit. When T 1 is switched on T 2 is driven and diode D 1 is forward-biased providing a reverse-bias keeping T 3 off. The base current I B is positive and saturates the power transistor T P . When T 1 is switched off,

T 3 switches on due to the negative path provided by R 3 , and

–V CC , providing a negative current for switching off the power T p transistor T P . When a negative power supply is not provided for the base

drive, a simple circuit like Fig. 3.16 can be used in low power

applications (step per motors, small dc–dc converters, relays,

pulsed circuits). When the input signal is high, T 1 switches

on and a positive current goes to T P keeping the capacitor charged with the zener voltage, when the input signal goes low

FIGURE 3.16 Base command without negative power supply.

36 M. G. Simoes

T R1

FIGURE 3.17 Antisaturation diodes (Baker’s clamp) improve power FIGURE 3.18 Isolated base drive circuit. transistor storage time.

T 2 provides a path for the discharge of the capacitor, imposing

a pulsed negative current from the base–emitter junction of T P .

A combination of large reverse base drive and antisaturation R 1 T

techniques may be used to reduce storage time to almost zero.

A circuit called Baker’s clamp may be employed as illustrated R 2 in Fig. 3.17. When the transistor is on, its base is two diode

drops below the input. Assuming that diodes D 2 and D 3 have a

forward-bias voltage of about 0.7 V, then the base will be 1.4 V

below the input terminal. Due to diode D 1 the collector is one

diode drop, or 0.7 V below the input. Therefore, the collector

will always be more positive than the base by 0.7 V, staying out of saturation, and because collector voltage increases, the

gain β also increases a little bit. Diode D 4 provides a negative

path for the reverse base current. The input base current can

be supplied by a driver circuit similar to the one discussed in FIGURE 3.19 Fig. 3.15. Transformer coupled base drive with tertiary winding

Several situations require ground isolation, off-line oper- transformer. ation, floating transistor topology, in addition safety needs

may call for an isolated base drive circuit. Numerous circuits have been demonstrated in switching power supplies isolated minimum number of components. The base transformer has topologies, usually integrating base drive requirements with

a tertiary winding which uses the energy stored in the trans- their power transformers. Isolated base drive circuits may pro- former to generate the reverse base current during the turn-off vide either constant current or proportional current excitation. command. Other configurations are also possible by adding to

A very popular base drive circuit for floating switching transis- the isolated circuits the Baker clamp diodes, or zener diodes tor is shown in Fig. 3.18. When a positive voltage is impressed with paralleled capacitors. on the secondary winding (V S ) of T R1

Sophisticated isolated base drive circuits can be used to pro- flows into the base of the power transistor T P which switches vide proportional base drive currents where it is possible to on (resistor R 1 limits the base current). The capacitor C 1 is control the value of β, keeping it constant for all collector cur- charged by (V S –V D1 –V BE ) and T 1 is kept blocked because the rents leading to shorter storage time. Figure 3.20 shows one diode D 1 reverse biases T 1 base–emitter. When V S is zipped of the possible ways to realize a proportional base drive cir- off, the capacitor voltage V C brings the emitter of T 1 to a neg- cuit. When transistor T 1 turns on, the transformer T R1 is in ative potential with respect to its base. Therefore, T 1 is excited negative saturation and the power transistor T P is off. Dur- so as to switch on and start pulling a reverse current from T P ing the time that T 1 is on, a current flows through winding base. Another very effective circuit is shown in Fig. 3.19 with a N 1 , limited by resistor R 1 , storing energy in the transformer,

a positive current

3 Power Bipolar Transistors 37

+V cc

FIGURE 3.20 Proportional base drive circuit.

holding it into saturation. When the transistor T 1 turns off, the energy stored in N 1 is transferred to winding N 4 , pulling

the core from negative to positive saturation. The windings N 2 and N 4 will withstand as a current source, the transistor T P will stay on and the gain β will be imposed by the turns ratio

given by Eq. (3.7). D 1

In order to use the proportional drive given in Fig. 3.20 careful design of the transformer must be done, so as to have flux balanced which will keep core under saturation. The transistor gain must be somewhat higher than the value imposed by

the transformer turns ratio, which requires cautious device matching.

The most critical portion of the switching cycle occurs during transistor turn-off, since normally reverse base cur- rent is made very large in order to minimize storage time, such conditions may avalanche the base–emitter junction lead-

FIGURE 3.21 ing to destruction. There are two options to prevent this Turn-off snubber network. from happening: turning off the transistor at low values of

collector–emitter voltage (which is not practical in most of the applications) or reducing collector current with rising col- the capacitor discharges through the resistor R back into the lector voltage, implemented by RC protective networks called transistor. snubbers. Therefore, an RC snubber network can be used to

It is not possible to fully develop all the aspects regard- divert the collector current during the turn-off improving ing simulation of BJT circuits. Before giving an example some the RBSOA in addition the snubber circuit dissipates a fair comments are necessary regarding modeling and simulation amount of switching power relieving the transistor. Figure 3.21 of BJT circuits. There are a variety of commercial circuit sim- shows a turn-off snubber network; when the power tran- ulation programs available on the market, extending from a

sistor is off, the capacitor C is charged through diode D 1 . set of functional elements (passive components, voltage con- Such collector current flows temporarily into the capacitor as trolled current sources, semiconductors) which can be used the collector-voltage rises; as the power transistor turns on, to model devices, to other programs with the possibility of

38 M. G. Simoes implementation of algorithm relationships. Those streams are outputs are to be printed or plotted. Two other statements are

called subcircuit (building auxiliary circuits around a SPICE required: the title statement and the end statement. The title primitive) and mathematical (deriving models from internal statement is the first line and can contain any information, device physics) methods. Simulators can solve circuit equa- while the end statement is always .END. This statement must tions exactly, given models for the non-linear transistors, and

be a line be itself, followed by a carriage return. In addition, predict the analog behavior of the node voltages and currents there are also comment statements, which must begin with an in continuous time. They are costly in computer time and such asterisk (*) and are ignored by SPICE. There are several model programs have not been written to usually serve the needs of equations for BJTs. designing power electronic circuits, rather for designing low-

SPICE has built-in models for the semiconductor devices, power and low-voltage electronic circuits. Therefore, one has and the user need to specify only the pertinent model to decide which approach should be taken for incorporating parameter values. The model for the BJT is based on the BJT power transistor modeling, and a trade-off between accu- integral-charge model of Gummel and Poon. However, if racy and simplicity must be considered. If precise transistor the Gummel–Poon parameters are not specified, the model modeling are required subcircuit oriented programs should reduces to piecewise-linear Ebers-Moll model as depicted in

be used. On the other hand, when simulation of complex Fig. 3.22. In either case, charge-storage effects, ohmic resis- power electronic system structures, or novel power electronic tances, and a current-dependent output conductance may be topologies are devised, the switch modeling should be rather included. The forward gain characteristics is defined by the

simple, by taking in consideration fundamental switching parameters I S and B F , the reverse characteristics by I S and B R . operations, and a mathematical oriented simulation program Three ohmic resistances R B ,R C , and R E are also included. The should be used.

two diodes are modeled by voltage sources, exponential equa- tions of Shockley can be transformed into logarithmic ones.

A set of device model parameters is defined on a separate .MODEL card and assigned a unique model name. The device

3.6 SPICE Simulation of Bipolar

element cards in SPICE then reference the model name. This

Junction Transistors

scheme lessens the need to specify all of the model parameters on each device element card. Parameter values are defined by

SPICE is a general-purpose circuit program that can be applied appending the parameter name, as given below for each model to simulate electronic and electrical circuits and predict the type, followed by an equal sign and the parameter value. Model circuit behavior. SPICE was originally developed at the Elec- parameters that are not given a value are assigned the default tronics Research Laboratory of the University of California, values given below for each model type. As an example, the Berkeley (1975), the name stands for: Simulation Program for Integrated Circuits Emphasis. A circuit must be specified in terms of element names, element values, nodes, variable

Collector

parameters, and sources. SPICE can do several types of circuit analyses:

i C • Non-linear dc analysis, calculating the dc transference.

R C • Non-linear transient analysis: calculates signals as a

function of time. • Linear ac analysis: computes a bode plot of output as a function of frequency.

C BC

F • Noise analysis.

R B i E • Sensitivity analysis.

Base

• Distortion analysis.

• Fourier analysis.

C BE

R • Monte-Carlo analysis.

i C In addition, PSpice has analog and digital libraries of stan-

dard components such as operational amplifiers, digital gates, flip-flops. This makes it a useful tool for a wide range of analog

and digital applications. An input file, called source file, con- E

sists of three parts: (1) data statements, with description of the components and the interconnections, (2) control statements, which tells SPICE what type of analysis to perform on the

Emitter

circuit, and (3) output statements, with specifications of what FIGURE 3.22 Ebers–Moll transistor model.

3 Power Bipolar Transistors 39

III

FIGURE 3.23 BJT buck chopper.

model parameters for the 2N2222A NPN transistor is given *ANALYSIS below:

.TRAN

2US 2.1MS

2MS UIC ;Transient analysis

;Graphics post-processor .MODEL Q2N2222A NPN (IS=14.34F XTI=3 EG=1.11

.PROBE

ABSTOL=1.OON RELTOL=0.01 VNTOL=0.1 VAF= 74.03 BF=255.9 NE=1.307 ISE=14.34F IKF=.2847

.OPTIONS

ITL5=40000 XTB=1.5 BR=6.092 NC=2 ISC=0 IKR=0 RC=1 CJC=7.306P

;Fourier analysis MJC=.3416 VJC=.75 FC=.5 CJE=22.01P MJE=.377 VJE=.75

.FOUR

25KHZ I (VY)

.END

TR=46.91N TF=411.1P ITF=.6 VTF=1.7 XTF=3 RB=10)

Figure 3.23 shows a BJT buck chopper. The dc input voltage

3.7 BJT Applications

145.84 µH, and the filter capacitance C is 200 µF. The chop- Bipolar junction power transistors are applied to a variety ping frequency is 25 kHz and the duty cycle of the chopper is of power electronic functions, switching mode power sup-

42% as indicated by the control voltage statement (V G ). The

listing below plots the instantaneous load current (I O ), the plies, dc motor inverters, PWM inverters just to name a few. input current (I S ), the diode voltage (V

D ), the output voltage

To conclude the present chapter, three applications are next

(V

C ), and calculate the Fourier coefficients of the input cur- rent (I ). It is suggested for the careful reader to have more

illustrated.

A flyback converter is exemplified in Fig. 3.24. The switching

details and enhancements on using SPICE for simulations on transistor is required to withstand the peak collector voltage specialized literature and references.

at turn-off and peak collector currents at turn-on. In order to limit the collector voltage to a safe value, the duty cycle must be

*SOURCE

kept relatively low, normally below 50%, i.e. 6. < 0.5. In prac-

VS 1 0 DC 12V

tice, the duty cycle is taken around 0.4, which limits the peak

VY 1 2 DC 0V ;Voltage source to measure

collector. A second design factor which the transistor must

input current

meet is the working collector current at turn-on, dependent

VG 7 3 PULSE 0V 30V 0.1NS 0/1Ns 16.7US 40US)

on the primary transformer-choke peak current, the primary-

*CIRCUIT

to-secondary turns ratio, and the output load current. When

RB 7 6 250

;Transistor base

the transistor turns on the primary current builds up in the

resistance

primary winding, storing energy, as the transistor turns off,

5 0 5 the diode at the secondary winding is forward biasing, releas-

L 3 4 145.8UH

ing such stored energy into the output capacitor and load. Such

C 5 0 200UF IC=3V

;Initial voltage

transformer operating as a coupled inductor is actually defined

VX 4 5 DC 0V ;Source to inductor

as a transformer-choke. The design of the transformer-choke

current

of the flyback converter must be done carefully to avoid sat-

DM 0 3 DMOD

;Freewheeling diode

uration because the operation is unidirectional on the B–H

.MODEL DMOD D(IS=2.22E–15 BV=1200V CJO=O TT=O)

characteristic curve.

Q1 2 6 3 3 2N6546 ;BJT switch

Therefore, a core with a relatively large volume and air gap

.MODEL 2N6546 NPN (IS=6.83E–14 BF=13 CJE=1PF

must be used. An advantage of the flyback circuit is the sim-

CJC=607.3PF TF=26.5NS)

plicity by which a multiple output switching power supply may

40 M. G. Simoes

FIGURE 3.24 Flyback converter.

TT FIGURE 3.25 Isolated forward converter.

be realized. This is because the isolation element acts as a com- the primary and secondary windings of the transformer have mon choke to all outputs, thus only a diode and a capacitor the same polarity, i.e. the dots are at the same winding ends. are needed for an extra output voltage.

When the transistor turns on, current builds up in the pri- Figure 3.25 shows the basic forward converter and its mary winding. Because of the same polarity of the transfo associated waveforms. The isolation element in the forward rmer secondary winding, such energy is forward transferred converter is a pure transformer which should not store energy, to the output and also stored in inductor L through diode D 2 and therefore, a second inductive element L is required at the which is forward-biased. When the transistor turns off, the output for proper and efficient energy transfer. Notice that transformer winding voltage reverses, back-biasing diode D 2

3 Power Bipolar Transistors 41

V CE I A I

L A , R A I A V CE V IN

T FIGURE 3.26 Chopper-fed dc drive.

and the flywheel diode D 3 is forward-biased, conducting also used in battery electric vehicles. A dc motor can be oper- current in the output loop and delivering energy to the load ated in one of the four quadrants by controlling the armature through inductor L. The tertiary winding and diode D, provide or field voltages (or currents). It is often required to reverse transformer demagnetization by returning the transformer the armature or field terminals in order to operate the motor magnetic energy into the output dc bus. It should be noted in the desired quadrant. Figure 3.26 shows a circuit arrange- that the duty cycle of the switch must be kept below 50%, ment of a chopper-fed dc separately excited motor. This is a so that when the transformer voltage is clamped through the one-quadrant drive, the waveforms for the armature voltage, tertiary winding, the integral of the volt-seconds between the load current, and input current are also shown. By varying the input voltage and the clamping level balances to zero. Duty duty cycle, the power flow to the motor (and speed) can be cycles above 50%, i.e. 6 > 0.5, will upset the volt-seconds bal- controlled. ance, driving the transformer into saturation, which in turn produces high collector current spikes that may destroy the switching transistor. Although the clamping action of the ter-

tiary winding and the diode limit the transistor peak collector Further Reading

voltage to twice the dc input, care must be taken during con- struction to couple the tertiary winding tightly to the primary

1. B. K. Bose, Power Electronics and Ac Drives, Prentice-Hall, Englewood (bifilar wound) to eliminate voltage spikes caused by leakage

Cliffs, NJ; 1986.

inductance. 2. G. C. Chryssis, High Frequency Switching Power Supplies: Theory and Chopper drives are connected between a fixed-voltage dc Design, McGraw-Hill, NY; 1984 3. N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics:

source and a dc motor to vary the armature voltage. In addition Converters, Applications, and Design, John Wiley & Sons, NY; 1995. to armature voltage control, a dc chopper can provide regener-

4. M. H. Rashid, Power Electronics: Circuits, Devices, and Applications, ative braking of the motors and can return energy back to the

Prentice-Hall, Englewood Cliffs, NJ; 1993. supply. This energy-saving feature is attractive to transporta-

5. B.W. Williams, Power Electronics: Devices, Drivers and Applications, tion systems as mass rapid transit (MRT), chopper drives are

John Wiley & Sons, NY; 1987.

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The Power MOSFET

Issa Batarseh, Ph.D.

4.1 Introduction .......................................................................................... 43

School of Electrical Engineering and Computer Science, University of

4.2 Switching in Power Electronic Circuits ........................................................ 44

Central Florida, 4000 Central

4.3 General Switching Characteristics .............................................................. 46

Florida Blvd., Orlando, Florida,

4.3.1 The Ideal Switch • 4.3.2 The Practical Switch

USA

4.4 The Power MOSFET................................................................................ 50

4.4.1 MOSFET Structure • 4.4.2 MOSFET Regions of Operation • 4.4.3 MOSFET Switching Characteristics • 4.4.4 MOSFET PSPICE Model • 4.4.5 MOSFET Large-signal Model • 4.4.6 Current MOSFET Performance

4.5 Future Trends in Power Devices ................................................................. 71 References ............................................................................................. 71

4.1 Introduction

performance. Because of their functional importance, drive complexity, fragility, cost, power electronic design engineer

In this chapter, an overview of power MOSFET (metal oxide must be equipped with the thorough understanding of the semiconductor field effect transistor) semiconductor switch- device operation, limitation, drawbacks, and related reliability ing devices will be given. The detailed discussion of the and efficiency issues. physical structure, fabrication, and physical behavior of the

In the 1980s, the development of power semiconductor device and packaging is beyond the scope of this chapter. devices took an important turn when new process technol- The emphasis here will be given on the terminal i–v switching ogy was developed that allowed the integration of MOS and characteristics of the available device, turn-on, and turn-off BJT technologies on the same chip. Thus far, two devices switching characteristics, PSPICE modeling and its current, using this new technology have been introduced: integrated voltage, and switching limits. Even though, most of today’s gate bipolar transistor (IGBT) and MOS controlled thyristor available semiconductor power devices are made of silicon (MCT). Many of the IC processing methods and equipment or germanium materials, other materials such as gallium have been adopted for the development of power devices. arsenide, diamond, and silicon carbide are currently being However, unlike microelectronic IC’s which process informa- tested.

tion, power devices IC’s process power, hence, their packaging One of the main contributions that led to the growth of the and processing techniques are quite different. Power semi- power electronics field has been the unprecedented advance- conductor devices represent the “heart” of modern power ment in the semiconductor technology, especially with respect electronics, with two major desirable characteristics of power to switching speed and power handling capabilities. The area semiconductor devices that guided their development are: the of power electronics started by the introduction of the sili- switching speed and power handling capabilities. con controlled rectifier (SCR) in 1958. Since then, the field

The improvement of semiconductor processing technol- has grown in parallel with the growth of the power semi- ogy along with manufacturing and packaging techniques has conductor device technology. In fact, the history of power allowed power semiconductor development for high voltage electronics is very much connected to the development of and high current ratings and fast turn-on and turn-off char- switching devices and it emerged as a separate discipline when acteristics. Today, switching devices are manufactured with high power bipolar junction transistors (BJTs) and MOSFETs amazing power handling capabilities and switching speeds devices where introduced in the 1960s and 1970s. Since as will be shown later. The availability of different devices then, the introduction of new devices has been accompanied with different switching speed, power handling capabilities, with dramatic improvement in power rating and switching size, cost, . . . etc. make it possible to cover many power

44 I. Batarseh electronics applications. As a result, trade-offs are made when

For more than forty years, it has been shown that in order it comes to selecting power devices.

to achieve high efficiency, switching (mechanical or electrical) is the best possible way to accomplish this. However, unlike mechanical switches, electronic switches are far more superior because of their speed and power handling capabilities as well as reliability.

4.2 Switching in Power Electronic

We should note that the advantages of using switches don’t

Circuits

come at no cost. Because of the nature of switch currents and voltages (square waveforms), normally high order harmon- ics are generated in the system. To reduce these harmonics,

As stated earlier, the heart of any power electronic circuit additional input and output filters are normally added to the is the semiconductor-switching network. The question arises system. Moreover, depending on the device type and power here is do we have to use switches to perform electrical power electronic circuit topology used, driver circuit control and conversion from the source to the load? The answer of course circuit protection can significantly increase the complexity of is no, there are many circuits which can perform energy con-

the system and its cost.

version without switches such as linear regulators and power amplifiers. However, the need for using semiconductor devices

E XAMPLE 4.1 The purpose of this example is to inves- to perform conversion functions is very much related to the

tigate the efficiency of four different power circuits converter efficiency. In power electronic circuits, the semicon-

whose functions are to take in power from 24 volts dc ductor devices are generally operated as switches, i.e. either in the on-state or the off-state. This is unlike the case in power

tive load. In other words, the tasks of these circuits are amplifiers and linear regulators where semiconductor devices

to serve as dc transformer with a ratio of 2:1. The four operate in the linear mode. As a result, very large amount of

circuits are shown in Fig. 4.1a–d representing voltage energy is lost within the power circuit before the processed

divider circuit, zener-regulator, transistor linear regula- energy reaches the output. The need to use semiconductor

tor, and switching circuit, respectively. The objective is switching devices in power electronic circuits is their ability to

to calculate the efficiency of those four power electronic control and manipulate very large amounts of power from the

circuits.

input to the output with a relatively very low power dissipa-

S OLUTION .

tion in the switching device. Hence, resulting in a very high efficient power electronic system.

Voltage Divider DC Regulator The first circuit is the Efficiency is considered as an important figure of merit and

simplest forming a voltage divider with R =R L has significant implications on the overall performance of the

and V o = 12 volts. The efficiency defined as the ratio system. Low efficient power systems means large amounts of

of the average load power, P L , to the average input power being dissipated in a form of heat, resulting in one or

power, P in

more of the following implications:

1. Cost of energy increases due to increased consumption.

2. Additional design complications might be imposed,

P in

especially regarding the design of device heat sinks.

= 50% size, and weight of the system, resulting in low power

3. Additional components such as heat sinks increase cost,

L +R

density.

4. High power dissipation forces the switch to operate In fact efficiency is simply V o /V in %. As the out- at low switching frequency, resulting in limited band-

put voltage becomes smaller, the efficiency decreases width, slow response, and most importantly, the size

proportionally.

and weight of magnetic components (inductors and transformers), and capacitors remain large. Therefore, it is always desired to operate switches at very high fre-

Zener DC Regulator Since the desired output is quencies. But, we will show later that as the switching

12 V, we select a zener diode with zener breakdown frequency increases, the average switching power dis-

V Z = 12 V. Assume the zener diode has the i–v char- sipation increases. Hence, a trade-off must be made

acteristic shown in Fig. 4.1e since R L between reduced size, weight, and cost of components

current, I L , is 2 A. Then we calculate R for I Z = 0.2 A vs reduced switching power dissipation, which means

(10% of the load current). This results in R inexpensive low switching frequency devices.

Since the input power is P in = 2.2 A × 24 V = 52.8 W

5. Reduced component and device reliability. and the output power is P out = 24 W. The efficiency of

4 The Power MOSFET 45

I L + V in

V in

(a)

(b)

(c)

(d)

(e)

v o V in V o,ave

0 DT

(f)

FIGURE 4.1 (a) Voltage divider; (b) zener regulator; (c) transistor regulator; (d) switching circuit; (e) i–v zener diode characteristics; and (f) switching waveform for S and corresponding output waveform.

46 I. Batarseh the circuit is given by,

power delivery with little output voltage ripple, then an LC filter must be added to smooth out the output

= voltage. %

24 W

The final observation is regarding what is known = 45.5% as load regulation and line regulation. The line reg- ulation is defined as the ratio between the change in output voltage, V o , with respect to the change in the

52.8 W

Transistor DC Regulator It is clear from Fig. 4.1c that input voltage V in . This is a very important param- for V o = 12 V, the collector emitter voltage must be

eter in power electronics since the dc input voltage around 12 V. Hence, the control circuit must provide

is obtained from a rectified line voltage that normally base current, I B to put the transistor in the active mode

changes by ±20%. Therefore, any off-line power elec- with V CE ≈ 12 V. Since the load current is 2 A, then col-

tronics circuit must have a limited or specified range

of line regulation. If we assume the input voltage in The total power dissipated in the transistor can be

lector current is approximately 2 A (assume small I B ).

Figs. 4.1a,b is changed by 2 V, i.e. V in = 2 V, with approximated by the following equation:

R L unchanged, the corresponding change in the output voltage V o is 1 V and 0.55 V, respectively. This is

P diss =V CE I C +V BE I B considered very poor line regulation. Figures 4.1c,d ≈V have much better line and load regulations since the

CE I C ≈ 12 × 2 = 24 watts

closed-loop control compensate for the line and load Therefore, the efficiency of the circuit is 50%.

variations.

Switching DC Regulator Let us consider the switch-

4.3 General Switching Characteristics

ing circuit of Fig. 4.1d by assuming the switch is ideal and periodically turned on and off is shown in Fig. 4.1f. The output voltage waveform is shown in Fig. 4.1f. Even

4.3.1 The Ideal Switch

though the output voltage is not constant or pure dc, its It is always desired to have the power switches perform as close average value is given by,

as possible to the ideal case. Device characteristically speaking, for a semiconductor device to operate as an ideal switch, it

1 must possess the following features:

V o,ave =

V in dt =V in D

1. No limit on the amount of current (known as forward

0 or reverse current) the device can carry when in the conduction state (on-state);

where D is the duty ratio equals the ratio of the on-time

2. No limit on the amount of the device-voltage ((known to the switching period. For V o,ave = 12 V, we set as forward or reverse blocking voltage) when the device

D = 0.5, i.e. the switch has a duty cycle of 0.5 or is in the non-conduction state (off-state); 50%. In case, the average output power is 48 W and

3. Zero on-state voltage drop when in the conduction the average input power is also 48 W, resulting in 100%

state;

efficiency! This is of course because we assumed the

4. Infinite off-state resistance, i.e. zero leakage current switch is ideal. However, let us assume a BJT switch when in the non-conduction state; and is used in the above circuit with V CE,sat = 1 V and I B

5. No limit on the operating speed of the device when is small, then the average power loss across the switch changes states, i.e. zero rise and fall times. is approximately 2 W, resulting in overall efficiency of

96%. Of course the switching circuit given in this exam- The switching waveforms for an ideal switch is shown in ple is over simplified, since the switch requires additional

Fig. 4.2, where i sw and v sw are the current through and the driving circuitry that was not shown, which also dis-

voltage across the switch, respectively. sipates some power. But still, the example illustrates

Both during the switching and conduction periods, the that high efficiency can be acquired by switching power

power loss is zero, resulting in a 100% efficiency, and with electronic circuits when compared to the efficiency of

no switching delays, an infinite operating frequency can be linear power electronic circuits. Also, the difference

achieved. In short, an ideal switch has infinite speed, unlimited between the linear circuit in Figs. 4.1b and c and the

power handling capabilities, and 100% efficiency. It must be switched circuit of Fig. 4.1d is that the power deliv-

noted that it is not surprising to find semiconductor-switching ered to the load in the later case in pulsating between

devices that can almost, for all practical purposes, perform as zero and 96 watts. If the application calls for constant

ideal switches for number of applications.

4 The Power MOSFET 47 i s

power gain, surge capacity, and over voltage capacity must w

be considered when addressing specific devices for specific applications. A useful plot that illustrates how switching takes

v sw

place from on to off and vice versa is what is called switch-

ing trajectory, which is simply a plot of i sw vs v sw . Figure 4.3b shows several switching trajectories for the ideal and practical

v sw

cases under resistive loads.

Consider a linear approximation of Fig. 4.3a as shown in Fig. 4.4a: (a) Give a possible V on

E XAMPLE 4.2

off

time

circuit implementation using a power switch whose switching waveforms are shown in Fig. 4.4a, (b) Derive the expressions for the instantaneous switching and

i sw conduction power losses and sketch them, (c) Determine I on

the total average power dissipated in the circuit during one switching frequency, and (d) The maximum power.

I off

time

S OLUTION . (a) First let us assume that the turn-on time, t on , and turn-off time, t off , the conduction voltage V ON ,

p (t ) and the leakage current, I OFF , are part of the switching characteristics of the switching device and have nothing to do with circuit topology.

When the switch is off, the blocking voltage across

time

the switch is V OFF that can be represented as a dc volt- age source of value V OFF reflected somehow across the

FIGURE 4.2 Ideal switching current, voltage, and power waveforms. switch during the off-state. When the switch is on, the

current through the switch equals I ON , hence, a dc cur- rent is needed in series with the switch when it is in the on-state. This suggests that when the switch turns

4.3.2 The Practical Switch off again, the current in series with the switch must

be diverted somewhere else (this process is known as The practical switch has the following switching and conduc-

commutation). As a result, a second switch is needed to tion characteristics:

carry the main current from the switch being investi- gated when it’s switched off. However, since i sw and v sw

1. Limited power handling capabilities, i.e. limited con- are linearly related as shown in Fig. 4.4b, a resistor will duction current when the switch is in the on-state, do the trick and a second switch is not needed. Figure 4.4 and limited blocking voltage when the switch is in the shows a one-switch implementation with S, the switch off-state. and R represents the switched-load.

2. Limited switching speed that is caused by the finite turn-on and turn-off times. This limits the maximum

(b) The instantaneous current and voltage waveforms operating frequency of the device.

during the transition and conduction times are given as

3. Finite on-state and off-state resistance’s i.e. there exists

follows

forward voltage drop when in the on-state, and reverse current flow (leakage) when in the off-state.

4. Because of characteristics 2 and 3 above, the practical

⎧ t ( I ⎪ ⎪ ⎪ t ON ON −I OFF ) +I OFF ⎪

0 ≤t≤t ON

switch experiences power losses in the on and the off

states (known as conduction loss), and during switch-

i sw (t ) =

I t ON ≤t≤T s −t ⎪ OFF ON

ing transitions (known as switching loss). Typical

− t −Ts tOFF ( I ON −I OFF ) +I OFF

T s −t OFF ≤t≤T s

switching waveforms of a practical switch are shown in Fig. 4.3a.

⎧ ⎪ V − OFF −V ⎪ ON ⎪

0 ≤t≤t ON

t ON

The average switching power and conduction power losses

× (t − t ON ) +V ON

can be evaluated from these waveforms. We should point out

v sw (t ) =

V ON

t ON ≤t≤T s −t OFF

the exact practical switching waveforms vary from one device

⎪ ⎪ ⎪ ⎪ ⎪ OFF −V to another device, but Fig. 4.3a is a reasonably good representa- ON ⎪ ⎪

T s −t OFF ≤t≤T s

t OFF

⎩ ⎪ tion. Moreover, other issues such as temperature dependence, ⎪ ×

s −t OFF )

ON

Forward voltage drop

V off

V on time

Turn-OFF i sw

delays I on

delays

I off time

Leakage current

p(t)

switching losses

P max

time

conduction losses

(a)

Typical practical

i sw

waveform

(Highly inductive load)

(Ideal switch) OFF OFF

ON (Resistive

load)

v sw

(b)

ON

OFF

V OFF

4 The Power MOSFET 49

(a)

(b)

(c)

FIGURE 4.4 Linear approximation of typical current and voltage switching waveforms.

It can be shown that if we assume I ON ≫I OFF and (c) The total average dissipated power is given by

V OFF ≫V ON , then the instantaneous power, p(t) = i sw v sw can be given as follows,

0 ≤t≤t ON

0 0 ON ⎨ ⎪ p(t ) = V ON I ON

ON

t ON ≤t≤T s −t OFF

T s −t OFF

⎪ ⎪ ⎪ ⎪ ⎪ − V OFF I ⎪ ON t 2 (t − (T s −t OFF )) (t −T s ) T s −t OFF ≤t≤T ⎩ s OFF

V ON I ON dt

t ON

s ⎤ Figure 4.4c shows a plot of the instantaneous power

V OFF I ON

− ⎥ −t OFF ))(t −T s )dt t ⎦

2 (t −(T s

where the maximum power during turn-on and off is

OFF

V OFF I ON /4.

s −t OFF

50 I. Batarseh The evaluation of the above integral gives

Drain(D)

The first expression represents the total switching

V GD V DS

(G) conduction loss over one switching cycle. We notice that −

loss, whereas the second expression represents the total

as the frequency increases, the average power increases

Gate(G)

linearly. Also the power dissipation increases with the

(S) increase in the forward conduction current and the V GS

Source(S)

reverse blocking voltage. (d) The maximum power occurs at the time when the

(b) first derivative of p(t) during switching is set to zero, i.e.

(a)

dp(t ) D D

dt t =t max

solve the above equation for t max , we obtain values at turn on and off, respectively,

t rise

t max = G G

2 t fall

t max =T−

S Solving for the maximum power, we obtain

(c)

(d)

V off I on

P max = FIGURE 4.5 Device symbols: (a) n-channel enhancement-mode;

4 (b) p-channel enhancement-mode; (c) n-channel depletion-mode; and (d) p-channel depletion-mode.